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Low Cost DC-500 MHz, 92 dB Logarithmic Amplifier AD8307
FEATURES
Complete multistage logarithmic amplifier 92 dB dynamic range: -75 dBm to +17 dBm to -90 dBm using matching network Single supply of 2.7 V minimum at 7.5 mA typ DC to 500 MHz operation, 1 dB linearity Slope of 25 mV/dB, intercept of -84 dBm Highly stable scaling over temperature Fully differential dc-coupled signal path 100 ns power-up time, 150 A sleep current
VPS 7
FUNCTIONAL BLOCK DIAGRAM
AD8307
7.5mA BAND GAP REFERENCE AND BIASING SIX 14.3dB 900MHz AMPLIFIER STAGES MIRROR 2A /dB 12.5k COM INPUT-OFFSET COMPENSATION LOOP
3 4 6
ENB INT
INP 8 INM 1
+INP 1.1k -INP 3
5
NINE DETECTOR CELLS SPACED 14.3dB COM 2
2
OUT
Conversion of signal level to decibel form Transmitter antenna power measurement Receiver signal strength indication (RSSI) Low cost radar and sonar signal processing Network and spectrum analyzers (to 120 dB) Signal level determination down to 20 Hz True decibel ac mode for multimeters
Figure 1.
GENERAL DESCRIPTION
The AD8307 is the first logarithmic amplifier made available in an 8-lead (SOIC-8) package. It is a complete 500 MHz monolithic demodulating logarithmic amplifier based on the progressive compression (successive detection) technique, providing a dynamic range of 92 dB to 3 dB law-conformance and 88 dB to a tight 1 dB error bound at all frequencies up to 100 MHz. It is extremely stable and easy to use, requiring no significant external components. A single-supply voltage of 2.7 V to 5.5 V at 7.5 mA is needed, corresponding to an unprecedented power consumption of only 22.5 mW at 3 V. A fast acting CMOScompatible control pin can disable the AD8307 to a standby current of less than 150 A. Each of the cascaded amplifier/limiter cells has a small signal gain of 14.3 dB, with a -3 dB bandwidth of 900 MHz. The input is fully differential and at a moderately high impedance (1.1 k in parallel with about 1.4 pF). The AD8307 provides a basic dynamic range extending from approximately -75 dBm (where dBm refers to a 50 source, that is, a sine amplitude of about 56 V) up to +17 dBm (a sine amplitude of 2.2 V). A simple input matching network can lower this range to -88 dBm to +3 dBm. The logarithmic linearity is typically within 0.3 dB up to 100 MHz over the central portion of this range, and degrades only slightly at 500 MHz. There is no minimum frequency limit. The AD8307 can be used at audio frequencies of 20 Hz or lower. The output is a voltage scaled 25 mV/dB, generated by a current of nominally 2 A/dB through an internal 12.5 k resistor. This voltage varies from 0.25 V at an input of -74 dBm (that is, the ac intercept is at -84 dBm, a 20 V rms sine input), up to 2.5 V for an input of +16 dBm. This slope and intercept can be trimmed using external adjustments. Using a 2.7 V supply, the output scaling can be lowered, for example to 15 mV/dB, to permit utilization of the full dynamic range. The AD8307 exhibits excellent supply insensitivity and temperature stability of the scaling parameters. The unique combination of low cost, small size, low power consumption, high accuracy and stability, very high dynamic range, and a frequency range encompassing audio through IF to UHF makes this product useful in numerous applications requiring the reduction of a signal to its decibel equivalent. The AD8307 operates over the industrial temperature range of -40C to +85C, and is available in 8-lead SOIC and PDIP packages.
Rev. C
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 (c)2006 Analog Devices, Inc. All rights reserved.
01082-001
APPLICATIONS
OFS
AD8307 TABLE OF CONTENTS
Features .............................................................................................. 1 Applications....................................................................................... 1 Functional Block Diagram .............................................................. 1 General Description ......................................................................... 1 Revision History ............................................................................... 2 Specifications..................................................................................... 3 Absolute Maximum Ratings............................................................ 4 ESD Caution.................................................................................. 4 Pin Configuration and Function Descriptions............................. 5 Typical Performance Characteristics ............................................. 6 Log Amp Theory .............................................................................. 9 Progressive Compression .......................................................... 10 Demodulating Log Amps .......................................................... 11 Intercept Calibration.................................................................. 12 Offset Control ............................................................................. 12 Extension of Range..................................................................... 13 Interfaces.......................................................................................... 14 Enable Interface .......................................................................... 14 Input Interface ............................................................................ 14 Offset Interface ........................................................................... 15 Output Interface ......................................................................... 15 Theory of Operation ...................................................................... 17 Basic Connections...................................................................... 17 Input Matching ........................................................................... 17 Narrow-Band Matching ............................................................ 18 Slope and Intercept Adjustments ............................................. 19 Applications Information .............................................................. 20 Buffered Output.......................................................................... 20 Four Pole Filter ........................................................................... 20 1 W to 1 kW 50 Power Meter............................................. 21 Measurement System with 120 dB Dynamic Range.............. 21 Operation at Low Frequencies.................................................. 22 DC-Coupled Applications......................................................... 22 Operation Above 500 MHz....................................................... 23 Outline Dimensions ....................................................................... 24 Ordering Guide .......................................................................... 24
REVISION HISTORY
10/06--Rev. B to Rev. C Updated Format..................................................................Universal Changes to Table 1............................................................................ 3 Changes to Table 3............................................................................ 5 Changes to Offset Interface........................................................... 15 Changes to Output Interface......................................................... 15 Updated captions to Outline Dimensions................................... 24 Changes to Ordering Guide .......................................................... 24 6/03--Rev. A to Rev. B Renumbered TPCs and Figures........................................Universal Changes to Ordering Guide ............................................................ 3 Changes to Figure 24...................................................................... 17 Deleted Evaluation Board Information ....................................... 18 Updated Outline Dimensions ....................................................... 19
Rev. C | Page 2 of 24
AD8307 SPECIFICATIONS
VS = 5 V, TA = 25C, RL 1 M, unless otherwise noted. Table 1.
Parameter GENERAL CHARACTERISTICS Input Range (3 dB Error) Input Range (1 dB Error) Logarithmic Conformance Logarithmic Slope vs. Temperature Logarithmic Intercept vs. Temperature Input Noise Spectral Density Operating Noise Floor Output Resistance Internal Load Capacitance Response Time Conditions From noise floor to maximum input From noise floor to maximum input f 100 MHz, central 80 dB f = 500 MHz, central 75 dB Unadjusted 1 Sine amplitude, unadjusted 2 Equivalent sine power in 50 Inputs shorted RSOURCE = 50 /2 Pin 4 to ground Small signal, 10% to 90%, 0 mV to100 mV, CL = 2 pF Large signal, 10% to 90%, 0 V to 2.4 V, CL = 2 pF AC-coupled input -3 dB Min Typ 92 88 0.3 0.5 25 20 -84 1.5 -78 12.5 3.5 400 500 500 10 900 14.3 3.2 1.6 50 0.8 1.1 1.4 10 Max Unit dB dB dB dB mV/dB mV/dB V dBm dBm nV/Hz dBm k pF ns ns MHz Hz MHz dB V V V V/C k pF A V mA A
1 27 27 -77 -76
23 23 -87 -88
10
15
Upper Usable Frequency 3 Lower Usable Frequency AMPLIFIER CELL CHARACTERISTICS Cell Bandwidth Cell Gain INPUT CHARACTERISTICS DC Common-Mode Voltage Common-Mode Range DC Input Offset Voltage 4 Incremental Input Resistance Input Capacitance Bias Current POWER INTERFACES Supply Voltage Supply Current Disabled
1 2 3
AC-coupled input Either input (small signal) RSOURCE 50 Drift Differential Either pin to ground Either input
-0.3
VS - 1 500
25 5.5 10 750
2.7 VENB 2 V VENB 1 V 8 150
This can be adjusted downward by adding a shunt resistor from the output to ground. A 50 k resistor reduces the nominal slope to 20 mV/dB. This can be adjusted in either direction by a voltage applied to Pin 5, with a scale factor of 8 dB/V. See the Operation Above 500 MHz section. 4 Normally nulled automatically by internal offset correction loop. May be manually nulled by a voltage applied between Pin 3 and ground; see the Applications Information section.
Rev. C | Page 3 of 24
AD8307 ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter Supply Input Voltage (Pin 1 and Pin 8) Storage Temperature Range, N, R Ambient Temperature Range, Rated Performance Industrial, AD8307AN, AD8307AR Lead Temperature Range (Soldering 10 sec) Ratings 7.5 V VSUPPLY -65C to +125C -40C to +85C
Stresses above those listed under Absolute Maximum Ratings can cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods can affect device reliability.
ESD CAUTION
300C
Rev. C | Page 4 of 24
AD8307 PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
INM 1 COM 2 OFS 3
8
INP VPS
01082-002
AD8307
7
6 ENB TOP VIEW OUT 4 (Not to Scale) 5 INT
Figure 2. Pin Configuration
Table 3. Pin Function Descriptions
Pin No. 1 2 3 4 5 6 7 8 Mnemonic INM COM OFS OUT INT ENB VPS INP Description Signal Input Minus Polarity. Normally at VPOS/2. Common Pin (Usually Grounded). Offset Adjustment. External capacitor connection. Logarithmic (RSSI) Output Voltage. ROUT = 12.5 k. Intercept Adjustment, 3 dB (see the Slope and Intercept Adjustments section). CMOS-Compatible Chip Enable. Active when high. Positive Supply: 2.7 V to 5.5 V. Signal Input Plus Polarity. Normally at VPOS/2. Due to the symmetrical nature of the response, there is no special significance to the sign of the two input pins. DC resistance from INP to INM = 1.1 k.
Rev. C | Page 5 of 24
AD8307 TYPICAL PERFORMANCE CHARACTERISTICS
8 7 2 6 5 4 3 2
01082-003
3
SUPPLY CURRENT (mA)
1
ERROR (dB)
TEMPERATURE ERROR @ +85C 0
-1
TEMPERATURE ERROR @ +25C TEMPERATURE ERROR @ -40C
0 1.0
1.1
1.2
1.3
1.4
1.5 VENB (V)
1.6
1.7
1.8
1.9
2.0
-3 -80
-60
-40
-20
0
20
INPUT LEVEL (dBm)
Figure 3. Supply Current vs. VENB Voltage (5 V)
8 7 6 5 4 3 2
Figure 6. Log Conformance vs. Input Level (dBm) at +25C, +85C, and -40C
3 INPUT FREQUENCY 10MHz
SUPPLY CURRENT (mA)
2
VOUT (V)
INPUT FREQUENCY 100MHz
1
INPUT FREQUENCY 300MHz
0 1.0
1.1
1.2
1.3
1.4
1.5 VENB (V)
1.6
1.7
1.8
1.9
2.0
0 -80
-60
-40
-20
0
20
INPUT LEVEL (dBm)
Figure 4. Supply Current vs. VENB Voltage (3 V)
3
Figure 7. VOUT vs. Input Level (dBm) at Various Frequencies
1.5
2
1.0
1
FREQUENCY INPUT = 300MHz ERROR (dB)
0.5 CFO VALUE = 0.01F 0
ERROR (dB)
0
-1 FREQUENCY INPUT = 100MHz -2
01082-005
-0.5
CFO VALUE = 1F CFO VALUE = 0.1F
-1.0
01082-008
-3 -80
-60
-40
-20
0
20
-1.5 -80
-60
-40
-20
0
20
INPUT LEVEL (dBm)
INPUT LEVEL (dBm)
Figure 5. Log Conformance vs. Input Level (dBm) 100 MHz, and 300 MHz
Figure 8. Log Conformance vs. CFO Values at 1 kHz Input Frequency
Rev. C | Page 6 of 24
01082-007
01082-004
1
INPUT FREQUENCY 500MHz
01082-006
1
-2
AD8307
3.0 INT PIN = 3.0V 10MHz, INT = -96.52dBm
3 100MHz 2
2.5
2.0
1
INT PIN = 4.0V 10MHz, INT = -87.71dBm
ERROR (dB)
VOUT (V)
+INPUT 0
1.5 NO CONNECT ON INT 10MHz, INT = -82.90dBm
1.0
-1
-INPUT
0.5
01082-009
-2
01082-012
0 -80
-70
-60
-50
-40
-30
-20
-10
0
10
20
-3 -80
-60
-40
-20
0
20
INPUT LEVEL (dBm)
INPUT LEVEL (dBm)
Figure 9. VOUT vs. Input Level at 5 V Supply; Showing Intercept Adjustment
3.0
Figure 12. Log Conformance vs. Input Level at 100 MHz Showing Response to Alternative Inputs
3
2.5
INT VOLTAGE INT = 1.0V, INT = -86dBm INT VOLTAGE INT NO CONNECT, INT = -71dBm
2
500MHz
2.0
1
1.5
ERROR (dB)
VOUT (V)
0
1.0
-1
100MHz
0.5
01082-010
0 -80
-70
-60
-50
-40
-30
-20
-10
0
10
-3 -90
-70
-50
-30
-10
10
INPUT LEVEL (dBm)
INPUT LEVEL (dBm)
Figure 10. VOUT vs. Input Level at 3 V Supply Using AD820 as Buffer, Gain = +2; Showing Intercept Adjustment
2.5
Figure 13. Log Conformance vs. Input at 100 MHz, 500 MHz; Input Driven Differentially Using Transformer
3
2.0
2
500MHz
1
VOUT (V)
1.5
100MHz @ -40C
ERROR (dB)
0 100MHz -1 10MHz
1.0
100MHz @ +25C
0.5
01082-011
0 -80
-60
-40
-20
0
20
-3 -70
-60
-50
-40
-30
-20
-10
0
10
20
INPUT LEVEL (dBm)
INPUT LEVEL (dBm)
Figure 11. VOUT vs. Input Level at Three Temperatures (-40C, +25C, +85C)
Figure 14. Log Conformance vs. Input Level at 3 V Supply Using AD820 as Buffer, Gain = +2
Rev. C | Page 7 of 24
01082-014
100MHz @ +85C
-2
01082-013
INT VOLTAGE INT = 2.0V, INT = -78dBm
-2
AD8307
CH1 200mV VOUT CH 1
2V CH1 500mV VOUT CH1
CH1 GND
VENB CH 2
INPUT SIGNAL CH2
CH2 GND
01082-015
CH2 2.00V
500ns
CH2 1.00V
200ns
Figure 15. Power-Up Response Time
CH1 200mV
2.5V CH1 500mV
Figure 18. VOUT Rise Time
VOUT CH 1
VENB CH 2
INPUT SIGNAL CH2
CH2 GND VOUT CH1
CH1 GND
01082-016
CH2 2.00V
500ns
CH2 1.00V
200ns
Figure 16. Power-Down Response Time
Figure 19. Large Signal Response Time
VPS = 5.0V HP8648B SIGNAL GENERATOR RF OUT 1nF NC
8 7 6 5
0.1F
HP8112A PULSE GENERATOR OUT
SYNCH OUT
HP8648B 10MHz REF CLK SIGNAL GENERATOR PULSE MODE IN PULSE VPS = 5.0V MODULATION MODE 0.1F 1nF RF OUT
8 7 6
EXT TRIG OUT
TRIG HP8112A OUT PULSE GENERATOR
NC
5
INP VPS ENB INT 52.3
1
INP VPS ENB INT 52.3
1
AD8307
INM COM OFS OUT
2 3 4
AD8307
INM COM OFS OUT
2 3 4
NC 1nF NC = NO CONNECT TEK P6139A 10x PROBE
01082-017
NC = NO CONNECT
Figure 17. Test Setup for Power-Up/Power-Down Response Time
Figure 20. Test Setup for VOUT Pulse Response
Rev. C | Page 8 of 24
01082-020
TEK744A SCOPE
NC
TRIG
1nF
TEK P6204 FET PROBE
TEK744A SCOPE
TRIG
01082-019
GND
01082-018
GND
AD8307 LOG AMP THEORY
Logarithmic amplifiers perform a more complex operation than that of classical linear amplifiers, and their circuitry is significantly different. A good grasp of what log amps do and how they work can prevent many pitfalls in their application. The essential purpose of a log amp is not to amplify, though amplification is utilized to achieve the function. Rather, it is to compress a signal of wide dynamic range to its decibel equivalent. It is thus a measurement device. A better term might be logarithmic converter, since its basic function is the conversion of a signal from one domain of representation to another, via a precise nonlinear transformation. Logarithmic compression leads to situations that can be confusing or paradoxical. For example, a voltage offset added to the output of a log amp is equivalent to a gain increase ahead of its input. In the usual case where all the variables are voltages, and regardless of the particular structure, the relationship between the variables can be expressed as:
VOUT = VY log (VIN /VX )
(the log intercept) at the unique value VIN = VX and ideally becomes negative for inputs below the intercept. In the ideal case, the straight line describing VOUT for all values of VIN continues indefinitely in both directions. The dotted line shows that the effect of adding an offset voltage VSHIFT to the output is to lower the effective intercept voltage VX. Exactly the same alteration could be achieved by raising the gain (or signal level) ahead of the log amp by the factor VSHIFT/VY. For example, if VY is 500 mV per decade (25 mV/dB), an offset of +150 mV added to the output appears to lower the intercept by two tenths of a decade, or 6 dB. Adding an offset to the output is thus indistinguishable from applying an input level that is 6 dB higher. The log amp function described by Equation 1 differs from that of a linear amplifier in that the incremental gain VOUT/VIN is a very strong function of the instantaneous value of VIN, as is apparent by calculating the derivative. For the case where the logarithmic base is ,
VOUT V =Y VIN VIN
(2)
(1)
where: VOUT is the output voltage. VY is the slope voltage; the logarithm is usually taken to base 10 (in which case VY is also the volts per decade). VIN is the input voltage. VX is the intercept voltage. All log amps implicitly require two references, here, VX and VY, which determine the scaling of the circuit. The absolute accuracy of a log amp cannot be any better than the accuracy of its scaling references. Equation 1 is mathematically incomplete in representing the behavior of a demodulating log amp such as the AD8307, where VIN has an alternating sign. However, the basic principles are unaffected, and this can be safely used as the starting point in the analyses of log amp scaling.
VOUT 5VY 4VY 3VY 2VY VY VOUT = 0 VIN = 10-2VX -40dBc -2V Y VIN = VX 0dBc VIN = 102VX +40dBc LOG VIN VIN = 104VX +80dBc
01082-021
That is, the incremental gain is inversely proportional to the instantaneous value of the input voltage. This remains true for any logarithmic base, which is chosen as 10 for all decibel related purposes. It follows that a perfect log amp is required to have infinite gain under classical small signal (zero amplitude) conditions. Less ideally, this result indicates that, whatever means are used to implement a log amp, accurate response under small signal conditions (that is, at the lower end of the dynamic range) demands the provision of a very high gain bandwidth product. A further consequence of this high gain is that, in the absence of an input signal, even very small amounts of thermal noise at the input of a log amp cause a finite output for zero input. This results in the response line curving away from the ideal shown in Figure 21 toward a finite baseline, which can be either above or below the intercept. Note that the value given for this intercept can be an extrapolated value, in which case the output can not cross zero, or even reach it, as is the case for the AD8307. While Equation 1 is fundamentally correct, a simpler formula is appropriate for specifying the calibration attributes of a log amp like the AD8307, which demodulates a sine wave input: VOUT = VSLOPE (PIN - P0) where: VOUT is the demodulated and filtered baseband (video or RSSI) output. VSLOPE is the logarithmic slope, now expressed in V/dB (typically between 15 mV/dB and 30 mV/dB). (3)
VSHIFT LOWER INTERCEPT
Figure 21. Ideal Log Amp Function
Figure 21 shows the input/output relationship of an ideal log amp, conforming to Equation 1. The horizontal scale is logarithmic and spans a wide dynamic range, shown here as over 120 dB, or six decades. The output passes through zero
PIN is the input power, expressed in decibels relative to some reference power level. P0 is the logarithmic intercept, expressed in decibels relative to the same reference level.
Rev. C | Page 9 of 24
AD8307
The most widely used reference in RF systems is decibels above 1 mW in 50 , written dBm. Note that the quantity (PIN - P0) is just dB. The logarithmic function disappears from the formula because the conversion has already been implicitly performed in stating the input in decibels. This is strictly a concession to popular convention; log amps manifestly do not respond to power (tacitly, power absorbed at the input), but rather to input voltage. The use of dBV (decibels with respect to 1 V rms) is more precise, though still incomplete, since waveform is involved, too. Since most users think about and specify RF signals in terms of power, more specifically, in dBm re: 50 , this convention is used in specifying the performance of the AD8307. in the case of the AD8307, VY is traceable to an on-chip band gap reference, while VX is derived from the thermal voltage kT/q and is later temperature corrected.
OUTPUT
AEK
A/1
SLOPE = 1
SLOPE = A
01082-023 01082-024
0
EK
INPUT
PROGRESSIVE COMPRESSION
Most high speed, high dynamic range log amps use a cascade of nonlinear amplifier cells (Figure 22) to generate the logarithmic function from a series of contiguous segments, a type of piecewise linear technique. This basic topology immediately opens up the possibility of enormous gain bandwidth products. For example, the AD8307 employs six cells in its main signal path, each having a small signal gain of 14.3 dB (x5.2) and a -3 dB bandwidth of about 900 MHz. The overall gain is about 20,000 (86 dB) and the overall bandwidth of the chain is some 500 MHz, resulting in the incredible gain bandwidth product (GBW) of 10,000 GHz, about a million times that of a typical op amp. This very high GBW is an essential prerequisite for accurate operation under small signal conditions and at high frequencies. In Equation 2, however, the incremental gain decreases rapidly as VIN increases. The AD8307 continues to exhibit an essentially logarithmic response down to inputs as small as 50 V at 500 MHz.
STAGE 1 STAGE 2 STAGE N-1 STAGE N
Figure 23. A/1 Amplifier Function
Let the input of an N-cell cascade be VIN, and the final output VOUT. For small signals, the overall gain is simply AN. A six stage system in which A = 5 (14 dB) has an overall gain of 15,625 (84 dB). The importance of a very high small signal gain in implementing the logarithmic function has been noted; however, this parameter is only of incidental interest in the design of log amps. From here onward, rather than considering gain, analyze the overall nonlinear behavior of the cascade in response to a simple dc input, corresponding to the VIN of Equation 1. For very small inputs, the output from the first cell is V1 = AVIN. The output from the second cell is V2 = A2 VIN, and so on, up to VN = AN VIN. At a certain value of VIN, the input to the Nth cell, VN-1, is exactly equal to the knee voltage EK. Thus, VOUT = AEK and since there are N-1 cells of gain A ahead of this node, calculate VIN = EK /AN-1. This unique situation corresponds to the lin-log transition, (labeled 1 in Figure 24). Below this input, the cascade of gain cells acts as a simple linear amplifier, while for higher values of VIN, it enters into a series of segments that lie on a logarithmic approximation (dotted line).
VOUT (4A-3) E K (3A-2) E K (A-1) EK (2A-1) E K AEK 0 EK/AN-1 EK/AN-2 EK/AN-3 EK/AN-4
1 2
VX
A
A
A
A
VW
01082-022
Figure 22. Cascade of Nonlinear Gain Cells
To develop the theory, first consider a scheme slightly different from that employed in the AD8307, but simpler to explain and mathematically more straightforward to analyze. This approach is based on a nonlinear amplifier unit, called an A/1 cell, with the transfer characteristic shown in Figure 23. The local small signal gain VOUT/VIN is A, maintained for all inputs up to the knee voltage EK, above which the incremental gain drops to unity. The function is symmetrical: the same drop in gain occurs for instantaneous values of VIN less than -EK. The large signal gain has a value of A for inputs in the range -EK VIN +EK, but falls asymptotically toward unity for very large inputs. In logarithmic amplifiers based on this amplifier function, both the slope voltage and the intercept voltage must be traceable to the one reference voltage, EK. Therefore, in this fundamental analysis, the calibration accuracy of the log amp is dependent solely on this voltage. In practice, it is possible to separate the basic references used to determine VY and VX and
2
3 3
RATIO OF A LOG VIN
Figure 24. First Three Transitions
Continuing this analysis, the next transition occurs when the input to the (N-1) stage just reaches EK; that is, when VIN = EK /AN-2. The output of this stage is then exactly AEK, and it is easily demonstrated (from the function shown in Figure 23) that the output of the final stage is (2A-1) EK (labeled 2 in Figure 24). Thus, the output has changed by an amount (A-1)EK for a change in VIN from EK /AN-1 to EK/AN-2, that is, a ratio change
Rev. C | Page 10 of 24
AD8307
of A. At the next critical point (labeled 3 in Figure 24), the input is again A times larger and VOUT has increased to (3A-2)EK, that is, by another linear increment of (A-1)EK.
OUTPUT
SLOPE = 0 AEK A/0 tanh
Further analysis shows that right up to the point where the input to the first cell is above the knee voltage, VOUT changes by (A-1)EK for a ratio change of A in VIN. This can be expressed as a certain fraction of a decade, which is simply log10(A). For example when A = 5, a transition in the piecewise linear output function occurs at regular intervals of 0.7 decade (log10(A), or 14 dB divided by 20 dB). This insight allows us to immediately write the volts per decade scaling parameter, which is also the scaling voltage, VY, when using base 10 logarithms, as
SLOPE = A
INPUT
Figure 25. A/0 Amplifier Functions (Ideal and Tanh)
VY =
Linear Change in VOUT Decades Change in VIN
=
( A - 1)EK
log10 ( A)
(4)
Note that only two design parameters are involved in determining VY, namely, the cell gain A and the knee voltage EK, while N, the number of stages, is unimportant in setting the slope of the overall function. For A = 5 and EK = 100 mV, the slope would be a rather awkward 572.3 mV per decade (28.6 mV/dB). A well designed log amp has rational scaling parameters. The intercept voltage can be determined by using two pairs of transition points on the output function (consider Figure 24). The result is
The AD640, AD606, AD608, AD8307, and various other Analog Devices, Inc. communications products incorporating a logarithmic IF amplifier all use this technique. It becomes apparent that the output of the last stage can no longer provide the logarithmic output, since this remains unchanged for all inputs above the limiting threshold, which occurs at VIN = EK/AN-1. Instead, the logarithmic output is now generated by summing the outputs of all the stages. The full analysis for this type of log amp is only slightly more complicated than that of the previous case. It is readily shown that, for practical purposes, the intercept voltage VX is identical to that given in Equation 5, while the slope voltage is
VY =
AEK log10 ( A )
01082-025
0
EK
(6)
VX =
A
EK ( N +1 / ( A -1))
(5)
For the case under consideration, using N = 6, calculate VZ = 4.28 V. However, be careful about the interpretation of this parameter, since it was earlier defined as the input voltage at which the output passes through zero (see Figure 21). Clearly, in the absence of noise and offsets, the output of the amplifier chain shown in Figure 23 can be zero when, and only when, VIN = 0. This anomaly is due to the finite gain of the cascaded amplifier, which results in a failure to maintain the logarithmic approximation below the lin-log transition (point 1 in Figure 24). Closer analysis shows that the voltage given by Equation 5 represents the extrapolated, rather than actual, intercept.
Preference for the A/0 style of log amp, over one using A/1 cells, stems from several considerations. The first is that an A/0 cell can be very simple. In the AD8307 it is based on a bipolar transistor differential pair, having resistive loads, RL, and an emitter current source, IE. This exhibits an equivalent knee voltage of EK = 2 kT/q and a small signal gain of A = IERL/EK. The large signal transfer function is the hyperbolic tangent (see dotted line in Figure 25). This function is very precise, and the deviation from an ideal A/0 form is not detrimental. In fact, the rounded shoulders of the tanh function result in a lower ripple in the logarithmic conformance than that obtained using an ideal A/0 function. An amplifier built of these cells is entirely differential in structure and can thus be rendered very insensitive to disturbances on the supply lines and, with careful design, to temperature variations. The output of each gain cell has an associated transconductance (gm) cell, which converts the differential output voltage of the cell to a pair of differential currents, which are summed simply by connecting the outputs of all the gm (detector) stages in parallel. The total current is then converted back to a voltage by a transresistance stage to generate the logarithmic output. This scheme is depicted, in single sided form, in Figure 26.
DEMODULATING LOG AMPS
Log amps based on a cascade of A/1 cells are useful in baseband applications because they do not demodulate their input signal. However, baseband and demodulating log amps alike can be made using a different type of amplifier stage, called an A/0 cell. Its function differs from that of the A/1 cell in that the gain above the knee voltage EK falls to zero, as shown by the solid line in Figure 25. This is also known as the limiter function, and a chain of N such cells are often used to generate hard limited output in recovering the signal in FM and PM modes.
Rev. C | Page 11 of 24
AD8307
AVIN VIN A/0 A/0 A2VIN A/0 A3VIN A/0 A4VIN VLIM
motion of VX resulting from the temperature variation of EK. Do this by adding an offset with the required temperature behavior. The precise temperature shaping of the intercept positioning offset results in a log amp having stable scaling parameters, making it a true measurement device, for example, as a calibrated received signal strength indicator (RSSI). In this application, one is more interested in the value of the output for an input waveform that is invariably sinusoidal. Although the input level can alternatively be stated as an equivalent power, in dBm, be sure to work carefully. It is essential to know the load impedance in which this power is presumed to be measured. In RF practice, it is generally safe to assume a reference impedance of 50 in which 0 dBm (1 mW) corresponds to a sinusoidal amplitude of 316.2 mV (223.6 mV rms). The intercept can likewise be specified in dBm. For the AD8307, it is positioned at -84 dBm, corresponding to a sine amplitude of 20 V. It is important to bear in mind that log amps do not respond to power, but to the voltage applied to their input. The AD8307 presents a nominal input impedance much higher than 50 (typically 1.1 k low frequencies). A simple input matching network can considerably improve the sensitivity of this type of log amp. This increases the voltage applied to the input and thus alters the intercept. For a 50 match, the voltage gain is 4.8 and the entire dynamic range moves down by 13.6 dB (see Figure 35). Note that the effective intercept is a function of waveform. For example, a square wave input reads 6 dB higher than a sine wave of the same amplitude and a Gaussian noise input 0.5 dB higher than a sine wave of the same rms value.
gm
gm
gm
gm
gm
01082-026
IOUT
Figure 26. Log Amp Using A/0 Stages and Auxiliary Summing Cells
The chief advantage of this approach is that the slope voltage can now be decoupled from the knee voltage EK = 2 kT/q, which is inherently PTAT. By contrast, the simple summation of the cell outputs would result in a very high temperature coefficient of the slope voltage given in Equation 6. To do this, the detector stages are biased with currents (not shown) which are rendered stable with temperature. These are derived either from the supply voltage (as in the AD606 and AD608) or from an internal band gap reference (as in the AD640 and AD8307). This topology affords complete control over the magnitude and temperature behavior of the logarithmic slope, decoupling it completely from EK. A further step is needed to achieve the demodulation response, required when the log amp is to convert an alternating input into a quasi-dc baseband output. This is achieved by altering the gm cells used for summation purposes to also implement the rectification function. Early discrete log amps based on the progressive compression technique used half-wave rectifiers. This made post-detection filtering difficult. The AD640 was the first commercial monolithic log amp to use a full wave rectifier, a practice followed in all subsequent Analog Devices types. These detectors can be modeled as being essentially linear gm cells, but producing an output current independent of the sign of the voltage applied to the input of each cell. That is, they implement the absolute value function. Since the output from the later A/0 stages closely approximates an amplitude symmetric square wave for even moderate input levels (most stages of the amplifier chain operate in a limiting mode), the current output from each detector is almost constant over each period of the input. Somewhat earlier detector stages produce a waveform having only very brief dropouts, while the detectors nearest the input produce a low level, almost sinusoidal waveform at twice the input frequency. These aspects of the detector system result in a signal that is easily filtered, resulting in low residual ripple on the output.
OFFSET CONTROL
In a monolithic log amp, direct coupling between the stages is used for several reasons. First, this avoids the use of coupling capacitors, which typically have a chip area equal to that of a basic gain cell, thus considerably increasing die size. Second, the capacitor values predetermine the lowest frequency at which the log amp can operate; for moderate values, this can be as high as 30 MHz, limiting the application range. Third, the parasitic (backplate) capacitance lowers the bandwidth of the cell, further limiting the applications. However, the very high dc gain of a direct-coupled amplifier raises a practical issue. An offset voltage in the early stages of the chain is indistinguishable from a real signal. For example, if it were as high as 400 V, it would be 18 dB larger than the smallest ac signal (50 V), potentially reducing the dynamic range by this amount. This problem is averted by using a global feedback path from the last stage to the first, which corrects this offset in a similar fashion to the dc negative feedback applied around an op amp. The high frequency components of the signal must be removed to prevent a reduction of the HF gain in the forward path. In the AD8307, this is achieved by an on-chip filter, providing sufficient suppression of HF feedback to allow operation above
INTERCEPT CALIBRATION
All monolithic log amps from Analog Devices include accurate means to position the intercept voltage VX (or equivalent power for a demodulating log amp). Using the scheme shown in Figure 26, the basic value of the intercept level departs considerably from that predicted by the simpler analyses given earlier. However, the intrinsic intercept voltage is still proportional to EK, which is PTAT (Equation 5). Recalling that the addition of an offset to the output produces an effect that is indistinguishable from a change in the position of the intercept, it is possible to cancel the left-right
Rev. C | Page 12 of 24
AD8307
1 MHz. To extend the range below this frequency, an external capacitor can be added. This permits the high-pass corner to be lowered to audio frequencies using a capacitor of modest value. Note that this capacitor has no effect on the minimum signal frequency for input levels above the offset voltage: this extends down to dc (for a signal applied directly to the input pins). The offset voltage varies from part to part; some exhibit essentially stable offsets of under 100 V without the benefit of an offset adjustment. applied to a tapped attenuator, and progressively smaller signals are applied to three passive rectifying gm cells whose outputs are summed with those of the main detectors. With care in design, the extension to the dynamic range can be seamless over the full frequency range. For the AD8307, it amounts to a further 27 dB. Therefore, the total dynamic range is theoretically 113 dB. The specified range of 90 dB (-74 dBm to +16 dBm) is for high accuracy and calibrated operation, and includes the low end degradation due to thermal noise and the top end reduction due to voltage limitations. The additional stages are not, however, redundant, but are needed to maintain accurate logarithmic conformance over the central region of the dynamic range, and in extending the usable range considerably beyond the specified range. In applications where log conformance is less demanding, the AD8307 can provide over 95 dB of range.
EXTENSION OF RANGE
The theoretical dynamic range for the basic log amp shown in Figure 26 is AN. For A = 5.2 (14.3 dB) and N = 6, it is 20,000 or 86 dB. The actual lower end of the dynamic range is largely determined by the thermal noise floor, measured at the input of the chain of amplifiers. The upper end of the range is extended upward by the addition of top end detectors. The input signal is
Rev. C | Page 13 of 24
AD8307 INTERFACES
The AD8307 comprises six main amplifier/limiter stages, each having a gain of 14.3 dB and small signal bandwidth of 900 MHz; the overall gain is 86 dB with a -3 dB bandwidth of 500 MHz. These six cells, and their associated gm styled full wave detectors, handle the lower two-thirds of the dynamic range. Three top end detectors, placed at 14.3 dB taps on a passive attenuator, handle the upper third of the 90 dB range. Biasing for these cells is provided by two references: one determines their gain; the other is a band gap circuit that determines the logarithmic slope and stabilizes it against supply and temperature variations. The AD8307 can be enabled/ disabled by a CMOS-compatible level at ENB (Pin 6). The first amplifier stage provides a low voltage noise spectral density (1.5 nV/Hz). The differential current-mode outputs of the nine detectors are summed and then converted to single sided form in the output stage, nominally scaled 2 A/dB. The logarithmic output voltage is developed by applying this current to an on-chip 12.5 k resistor, resulting in a logarithmic slope of 25 mV/dB (that is, 500 mV/decade) at Pin OUT. This voltage is not buffered, allowing the use of a variety of special output interfaces, including the addition of post-demodulation filtering. The last detector stage includes a modification to temperature stabilize the log intercept, which is accurately positioned to make optimal use of the full output voltage range available. The intercept can be adjusted using the INT pin, which adds or subtracts a small current to the signal current.
AD8307
VPS 7 7.5mA BAND GAP REFERENCE AND BIASING SIX 14.3dB 900MHz AMPLIFIER STAGES MIRROR 2A /dB 12.5k COM
3
01082-027
tolerance is typically within 20%. Similarly, the capacitors have a typical tolerance of 15% and essentially zero temperature or voltage sensitivity. Most interfaces have additional small junction capacitances associated with them, due to active devices or ESD protection; these can be neither accurate nor stable. Component numbering in each of these interface diagrams is local.
ENABLE INTERFACE
The chip enable interface is shown in Figure 28. The currents in the diode-connected transistors control the turn on and turn off states of the band gap reference and the bias generator, and are a maximum of 100 A when Pin 6 is taken to 5 V, under worstcase conditions. Left unconnected, or at a voltage below 1 V, the AD8307 is disabled and consume a sleep current of under 50 A; tied to the supply, or a voltage above 2 V, it is fully enabled. The internal bias circuitry is very fast, typically <100 ns for either off or on. In practice, the latency period before the log amp exhibits its full dynamic range is more likely to be limited by factors relating to the use of ac coupling at the input or the settling of the offset control loop.
40k ENB 6
AD8307
TO BIAS STAGES
01082-028
COM
2
Figure 28. Enable Interface
7
VPS
S
6
ENB
COM
125
125
6k 2k 2k 6k Q1
INP 8 INM 1
+INP 1.1k -INP 3
5
INT
INP
8
CP
NINE DETECTOR CELLS SPACED 14.3dB COM 2 INPUT-OFFSET COMPENSATION LOOP
2
4
OUT
INM
1
CD
4k
TOP-END DETECTORS TYP 2.2V FOR 3V SUPPLY, 3.2V AT 5V
~3k Q2
CM
OFS
Figure 27. Main Features of the AD8307
2
COM
The last gain stage also includes an offset sensing cell. This generates a bipolarity output current when the main signal path has an imbalance due to accumulated dc offsets. This current is integrated by an on-chip capacitor (which can be increased in value by an off-chip component at OFS). The resulting voltage is used to null the offset at the output of the first stage. Since it does not involve the signal input connections, whose ac-coupling capacitors otherwise introduce a second pole in the feedback path, the stability of the offset correction loop is assured. The AD8307 is built on an advanced, dielectrically isolated, complementary bipolar process. Most resistors are thin film types having a low temperature coefficient of resistance (TCR) and high linearity under large signal conditions. Their absolute
Figure 29. Signal Input Interface
INPUT INTERFACE
Figure 29 shows the essentials of the signal input interface. CP and CM are the parasitic capacitances to ground; CD is the differential input capacitance, mostly due to Q1 and Q2. In most applications, both input pins are ac-coupled. The switches close when Enable is asserted. When disabled, the inputs float, bias current IE is shut off, and the coupling capacitors remain charged. If the log amp is disabled for long periods, small leakage currents discharge these capacitors. If they are poorly matched, charging currents at power-up can generate a
Rev. C | Page 14 of 24
01082-029
COM
S
IE 2.4mA
AD8307
transient input voltage that can block the lower reaches of the dynamic range until it has become much less than the signal. In most applications, the signal is single sided and can be applied to either Pin 1 or Pin 8, with the other pin ac-coupled to ground. Under these conditions, the largest input signal that can be handled by the AD8307 is 10 dBm (sine amplitude of 1 V) when operating from a 3 V supply; 16 dBm can be handled using a 5 V supply. The full 16 dBm can be achieved for supplies down to 2.7 V, using a fully balanced drive. For frequencies above about 10 MHz, this is most easily achieved using a matching network. Using such a network, having an inductor at the input, the input transient is eliminated. Occasionally, it is desirable to use the dc-coupled potential of the AD8307. The main challenge here is to present signals to the log amp at the elevated common-mode input level, requiring the use of low noise, low offset buffer amplifiers. Using dual supplies of 3 V, the input pins can operate at ground potential.
7
VPS
INPUT STAGE
125
125 MAIN GAIN STAGES Q2 64A AT BALANCE S TO LAST DETECTOR
Q1
gm
AVERAGE ERROR CURRENT CHP
2
01082-030
BIAS, ~1.2V
OFS Q3 36k Q4 48k
3
COFS
COM
Figure 30. Offset Interface and Offset Nulling Path
The offset feedback is limited to a range of 1.6 mV; signals larger than this override the offset control loop, which only affects performance for very small inputs. An external capacitor reduces the high-pass corner to arbitrarily low frequencies; using 1 F this corner is below 10 Hz. All ADI log amps use an offset nulling loop; the AD8307 differs in using this single sided form.
OUTPUT INTERFACE
The outputs from the nine detectors are differential currents, having an average value that is dependent on the signal input level, plus a fluctuation at twice the input frequency. The currents are summed at nodes LGP and LGM in Figure 31. Further currents are added at these nodes, to position the intercept, by slightly raising the output for zero input, and to provide temperature compensation. Since the AD8307 is not laser trimmed, there is a small uncertainty in both the log slope and the log intercept. These scaling parameters can be adjusted. For zero signal conditions, all the detector output currents are equal. For a finite input of either polarity, their difference is converted by the output interface to a single sided unipolar current nominally scaled 2 A/dB (40 A/decade), at Pin OUT. An on-chip 12.5 k resistor, R1, converts this current to a voltage of 25 mV/dB. C1 and C2 are effectively in shunt with R1 and form a low-pass filter pole with a corner frequency of about 5 MHz. The pulse response settles to within 1% of the final value within 300 ns. This integral low-pass filter provides adequate smoothing in many IF applications. At 10.7 MHz, the 2f ripple is 12.5 mV in amplitude, equivalent to 0.5 dB, and only 0.5 mV (0.02 dB) at f = 50 MHz. A filter capacitor CFLT added from Pin OUT to ground lowers this corner frequency. Using 1 F, the ripple is maintained to less than 0.5 dB down to input frequencies of 100 Hz. Note that COFS should also be increased in low frequency applications, and is typically made equal to CFLT.
OFFSET INTERFACE
The input referred dc offsets in the signal path are nulled via the interface associated with Pin 3, shown in Figure 30. Q1 and Q2 are the first stage input transistors, with their corresponding load resistors (125 ). Q3 and Q4 generate small currents, which can introduce a dc offset into the signal path. When the voltage on OFS is at about 1.5 V, these currents are equal, and nominally 64 A. When OFS is taken to ground, Q4 is off and the effect of the current in Q3 is to generate an offset voltage of 64 V x 125 = 8 mV. Since the first stage gain is x5, this is equivalent to a input offset (INP to INM) of 1.6 mV. When OFS is taken to its most positive value, the input referred offset is reversed to -1.6 mV. If true dc coupling is needed, down to very small inputs, this automatic loop must be disabled, and the residual offset eliminated using a manual adjustment. In normal operation, however, using an ac-coupled input signal, the OFS pin should be left open. Any residual input offset voltage is then automatically nulled by the action of the feedback loop. The gm cell, which is gated off when the chip is disabled, converts any output offset (sensed at a point near the end of the cascade of amplifiers) to a current. This is integrated by the on-chip capacitor CHP, and any added external capacitance COFS, so as to generate an error voltage, which is applied back to the input stage in the polarity needed to null the output offset. From a small signal perspective, this feedback alters the response of the amplifier, which, rather than behaving as a fully dc-coupled system, now exhibits a zero in its ac transfer function, resulting in a closed loop high-pass corner at about 1.5 MHz.
Rev. C | Page 15 of 24
AD8307
It can be desirable to increase the speed of the output response, with the penalty of increased ripple. One way to do this is simply by connecting a shunt load resistor from Pin OUT to ground, which raises the low-pass corner frequency. This also alters the logarithmic slope, for example to 7.5 mV/dB using a 5.36 k resistor, while reducing the 10% to 90% rise time to 25 ns. The ripple amplitude for 50 MHz input remains 0.5 mV, but this is now equivalent to 0.07 dB. If a negative supply is available, the output pin can be connected directly to the summing node of an external op amp connected as an inverting mode transresistance stage. Note that while the AD8307 can operate down to supply voltages of 2.7 V, the output voltage limit is reduced when the supply drops below 4 V. This characteristic is the result of necessary headroom requirements, approximately two VBE drops, in the design of the output stage.
7
VPS
3pF LGP LGM FROM ALL DETECTORS 2A/dB 0-220A 25mV/dB OUT 4 CFLT C1 2.5pF
1.25k
1.25k ~400mV
8.25k
60k 1.25k 1.25k
5
INT
C2 1pF R1 12.5k
BIAS 60A
2
01082-031
COM
Figure 31. Simplified Output Interface
Rev. C | Page 16 of 24
AD8307 THEORY OF OPERATION
The AD8307 has very high gain and a bandwidth from dc to over 1 GHz, at which frequency the gain of the main path is still over 60 dB. Consequently, it is susceptible to all signals within this very broad frequency range that find their way to the input terminals. It is important to remember that these are indistinguishable from the wanted signal, and has the effect of raising the apparent noise floor (that is, lowering the useful dynamic range). For example, while the signal of interest can be an IF of 50 MHz, any of the following could easily be larger than the IF signal at the lower extremities of its dynamic range: 60 Hz hum (picked up due to poor grounding techniques); spurious coupling (from a digital clock source on the same PC board); local radio stations; and so on. Careful shielding is essential. A ground plane should be used to provide a low impedance connection to the common pin, COM, for the decoupling capacitor(s) used at VPS, and as the output ground. It is inadvisable to assume that the ground plane is an equipotential. Neither of the inputs should be ac-coupled directly to the ground plane, but should be kept separate from it, being returned instead to the low associated with the source. This can mean isolating the low side of an input connector with a small resistance to the ground plane. the log amp side of the coupling capacitors; in the former case, smaller capacitors can be used for a given frequency range; in the latter case, the effective RIN is lowered directly at the log amp inputs. Figure 33 shows the output versus the input level, in dBm, when driven from a terminated 50 generator, for sine inputs at 10 MHz, 100 MHz, and 500 MHz; Figure 34 shows the typical logarithmic conformance under the same conditions. Note that +10 dBm corresponds to a sine amplitude of 1 V, equivalent to an rms power of 10 mW in a 50 termination. However, if the termination resistor is omitted, the input power is negligible. The use of dBm to define input level therefore needs to be considered carefully in connection with the AD8307.
3.0
2.5 10MHz
OUTPUT VOLTAGE (V)
2.0
1.5 100MHz 1.0 500MHz
01082-033
BASIC CONNECTIONS
Figure 32 shows the simple connections suitable for many applications. The inputs are ac coupled by C1 and C2, which should have the same value, say, CC. The coupling time constant is RIN CC/2, thus forming a high-pass corner with a 3 dB attenuation at fHP = 1/(pRINCC ). In high frequency applications, fHP should be as large as possible in order to minimize the coupling of unwanted low frequency signals. Conversely, in low frequency applications, a simple RC network forming a lowpass filter should be added at the input for the same reason. For the case where the generator is not terminated, the signal range should be expressed in terms of the voltage response, and extends from -85 dBV to +6 dBV.
0.1F 4.7 VP, 2.7V TO 5.5V AT ~8mA
0.5
0 -80
-70
-60
-50
-40
-30
-20
-10
0
10
20
INPUT LEVEL (dBm)
Figure 33. Log Response at 10 MHz, 100 MHz, and 500 MHz
5 4 3 2
ERROR (dB)
500MHz
1 0 -1 -2 -3 10MHz 100MHz
C1 = CC NC INPUT -75dBm TO +16dBm RT
8 7 6 5
-4 -5 -80 -70 -60 -50 -40 -30 -20 -10 0 10 20
RIN 1.1k
INP VPS ENB INT
AD8307
INM COM OFS OUT
1 2 3 4
INPUT LEVEL (dBm)
Figure 34. Logarithmic Law Conformance at 10 MHz, 100 MHz, and 500 MHz
OUTPUT 25mV/dB
01082-032
NC C2 = CC NC = NO CONNECT
INPUT MATCHING
Where higher sensitivity is required, an input matching network is valuable. Using a transformer to achieve the impedance transformation also eliminates the need for coupling capacitors, which lowers the offset voltage generated directly at the input, and balances the drives to Pin INP and Pin INM. The choice of turns ratio depends somewhat on the frequency. At frequencies below 50 MHz, the reactance of the input capacitance is much higher than the real part of the input impedance. In this
Figure 32. Basic Connections
Where it is necessary to terminate the source at a low impedance, the resistor RT should be added, with allowance for the shunting effect of the basic 1.1 k input resistance (RIN) of the AD8307. For example, to terminate a 50 source, a 52.3 1% tolerance resistor should be used. This can be placed on the input side or
Rev. C | Page 17 of 24
01082-034
AD8307
frequency range, a turns ratio of about 1:4.8 lowers the input impedance to 50 while raising the input voltage, thus lowering the effect of the short-circuit noise voltage by the same factor. There is a small contribution from the input noise current, so the total noise is reduced by a lesser factor. The intercept is also lowered by the turns ratio; for a 50 match, it is reduced by 20 log10 (4.8) or 13.6 dB.
0.1F 4.7 VP, 2.7V TO 5.5V AT ~8mA C1 NC 50 INPUT -88dBm TO +3dBm
8 7 6 5
INP VPS ENB INT LM ZIN = 50 C2 NC = NO CONNECT
1
AD8307
INM COM OFS OUT
2 3 4
Transformer coupling is useful in broadband applications. However, a magnetically-coupled transformer may not be convenient in some situations. At high frequencies, it is often preferable to use a narrow-band matching network, as shown in Figure 35. This has several advantages. The same voltage gain is achieved, providing increased sensitivity, but now a measure of selectivity is also introduced. The component count is low: two capacitors and an inexpensive chip inductor. Further, by making these capacitors unequal, the amplitudes at Pin INP and Pin INM can be equalized when driving from a single sided source; that is, the network also serves as a balun. Figure 36 shows the response for a center frequency of 100 MHz. Note the very high attenuation at low frequencies. The high frequency attenuation is due to the input capacitance of the log amp.
Figure 35. High Frequency Input Matching Network
14 13 12 11 10 9 8 7 6 5 4 3 2 1 0 -1 60 70 80 90 100 110 120 130 140 INPUT
01082-036
GAIN
DECIBELS
01082-035
NARROW-BAND MATCHING
NC
OUTPUT 25mV/dB
150
FREQUENCY (MHz)
Figure 36. Response of 100 MHz Matching Network
Table 4 provides solutions for a variety of center frequencies (FC) and matching impedances (ZIN) of nominally 50 and 100 . The unequal capacitor values were chosen to provide a well balanced differential drive, and to allow better centering of the frequency response peak when using standard value components; this generally results in a ZIN that is not exact. The full AD8307 HF input impedance and the inductor losses are included in the modeling.
Table 4. Narrow-Band Matching Values
FC (MHz) 10 20 50 100 150 200 250 500 10 20 50 100 150 200 250 500 ZIN () 45 44 46 50 57 57 50 54 103 102 99 98 101 95 92 114 C1 (pF) 160 82 30 15 10 7.5 6.2 3.9 100 51 22 11 7.5 5.6 4.3 2.2 C2 (pF) 150 75 27 13 8.2 6.8 5.6 3.3 91 43 18 9.1 6.2 4.7 3.9 2.0 LM (nH) 3300 1600 680 330 220 150 100 39 5600 2700 1000 430 260 180 130 47 Voltage Gain (dB) 13.3 13.4 13.4 13.4 13.2 12.8 12.3 10.9 10.4 10.4 10.6 10.5 10.3 10.3 9.9 6.8
Rev. C | Page 18 of 24
AD8307
SLOPE AND INTERCEPT ADJUSTMENTS
Where higher calibration accuracy is needed, the adjustments shown in Figure 37 can be used, either singly or in combination. The log slope is lowered to 20 mV/dB by shunting the nominally 12.5 k on-chip load resistor (see Figure 31) with 50 k, adjusted by VR1. The calibration range is 10% (18 mV/dB to 22 mV/dB), including full allowance for the variability in the value of the internal load. The adjustment can be made by alternately applying two input levels, provided by an accurate signal generator, spaced over the central portion of the log amp's dynamic range, for example -60 dBm and 0 dBm. An AM modulated signal, at the center of the dynamic range, can also be used. For a modulation depth M, expressed as a fraction, the decibel range between the peaks and troughs over one cycle of the modulation period is given by The log intercept is adjustable over a 3 dB range, which is sufficient to absorb the worst-case intercept error in the AD8307, plus some system level errors. For greater range, set RS to zero. VR2 is adjusted while applying an accurately known CW signal near the lower end of the dynamic range in order to minimize the effect of any residual uncertainty in the slope. For example, to position the intercept to -80 dBm, a test level of -65 dBm can be applied and VR2 adjusted to produce a dc output of 15 dB above zero at 25 mV/dB, which is +0.3 V.
0.1F 4.7 VR2 50k VP, 2.7V TO 5.5V AT ~8mA RS
C1 = CC
8 7 6
3dB
5
dB = 20 log10
1+ M 1- M
(7)
INPUT -75dBm TO +16dBm
INP VPS ENB INT
AD8307
INM COM OFS OUT
1 2 3 4
FOR VP = 3V, RS = 20k VP = 5V, RS = 51k
For example, using an rms signal level of -40 dBm with a 70% modulation depth (M = 0.7), the decibel range is 15 dB, as the signal varies from -47.5 dBm to -32.5 dBm.
NC C2 = CC NC = NO CONNECT 32.4k VR1 50k
Figure 37. Slope and Intercept Adjustments
Rev. C | Page 19 of 24
01082-037
20mV/dB 10%
AD8307 APPLICATIONS INFORMATION
The AD8307 is a highly versatile and easily applied log amp requiring very few external components. Most applications of this product can be accommodated using the simple connections shown in the preceding section.
INPUT -75dBm TO +16dBm 0.1F 4.7 VR2 50k RS 2N3904 VP, 2.7V TO 5.5V
3dB
8 7 6 5
INP VPS ENB INT
AD8307
INM COM OFS OUT
1 2 3 4
AD8031
10mV/dB 18%
25mV/dB R2 3.01k RT (OPTIONAL) OUTPUT 50 MINIMUM
01082-039
01082-040
BUFFERED OUTPUT
The output can be buffered, and the slope optionally increased, using an op amp. If the single-supply capability is to be preserved, a suitable component is the AD8031. Like the AD8307, it is capable of operating from a 2.7 V supply and features a rail-torail output capability; it is available in a 5-lead version and in dual form as the 8-lead AD8032. Figure 38 shows how the slope can be increased to 50 mV/dB (1 V per decade), requiring a 5 V supply (90 dB times 50 mV is a 4.5 V swing). VR1 provides a 10% slope adjustment; VR2 provides a 3 dB intercept range. With R2 = 4.99 k, the slope is adjustable to 25 mV/dB, allowing the use of a 2.7 V supply. Setting R2 to 80.6 k, it is raised to 100 mV/dB, providing direct reading in decibels on a digital voltmeter. Since a 90 dB range now corresponds to a 9 V swing, a supply of at least this amount is needed for the op amp.
0.1F 4.7 VR2 50k VP, 2.7V TO 5.5V RS FOR VP = 3V, RS = 20k VP = 5V, RS = 51k
NC NC = NO CONNECT
6.34k VR1 5k
R1 2k COM
Figure 39. Cable Driving Log Amp
In Figure 40, the capacitor values are chosen for operation in the audio field, providing a corner frequency of 10 Hz, an attenuation of 80 dB/decade above this frequency, and a 1% settling time of 150 ms (0.1% in 175 ms). The residual ripple is 4 mV (0.02 dB) when the input to the AD8307 is at 20 Hz. This filter can easily be adapted to other frequencies by proportional scaling of C5 to C7 (for example, for 100 kHz use 100 pF). Placed ahead of a digital multimeter, the convenient slope scaling of 100 mV/dB requires only a repositioning of the decimal point to read directly in decibels. The supply voltage for the filter must be large enough to support the dynamic range; a minimum of 9 V is needed for most applications; 12 V is recommended.
INPUT 5mV TO 160V rms 0.1F R1 50k C1 10F
8
3dB
8 7 6 5
INPUT -75dBm TO +16dBm
INP VPS ENB INT
AD8307
INM COM OFS OUT
1 2 3 4
AD8031
OUTPUT 50mV/dB 10% R2 30.1k
01082-038
4.7 VR1 2k
VP 422 C5 1F +
OP AMP IS AD8032 SCALE C1 TO C8 AS NEEDED. NOTE POLARITIES IF TANTALUM CAPACITORS ARE USED. C8 7.32k 1F + 100k 34k 34k C7 + 1F 80.6k COM 93k OUTPUT 100mV/dB
NC NC = NO CONNECT
20mV/dB
+
7
INT 4dB NC
32.4k VR1 50k
C1
R1 20k COM
6
5
INP VPS ENB INT C3 2.5nF
AD8307
INM COM OFS OUT
1 2 3 4
Figure 38. Log Amp with Buffered Output
C1 is optional; it lowers the corner frequency of the low-pass output filter. A value of 0.1 F should be used for applications in which the output is measured on a voltmeter or other low speed device. On the other hand, when C1 is omitted, the 10% to 90% response time is under 200 ns and is typically 300 ns to 99% of final value. To achieve faster response times, it is necessary to lower the load resistance at the output of the AD8307, then restore the scale using a higher gain in the op amp. Using 8.33 k, the basic slope is 10 mV/dB; this can be restored to 25 mV/dB using a buffer gain of 2.5. The overall 10% to 90% response time is under 100 ns. Figure 39 shows how the output current capability can be augmented to drive a 50 load; RT optionally provides reverse termination, which halves the slope to 12.5 mV/dB.
+
+ C2 10F
VR2 50k SLOPE C6 + 32.4k 1F
C4 1F
75k
Figure 40. Log Amp with Four Pole Low Pass Filter
FOUR POLE FILTER
In low frequency applications, for example, audio down to 20 Hz, it is useful to employ the buffer amplifier as a multipole low-pass filter in order to achieve low output ripple while maintaining a rapid response time to changes in signal level.
Figure 40 also shows the use of an input attenuator that can optionally be employed here, or in any other of these applications, to produce a useful wide range ac voltmeter with direct decibel scaling. The basic range of -73 dBm to +17 dBm (that is, 50 V rms to 1.6 V rms, for sine excitations) is shifted for illustrative purposes to 5 mV to 160 V rms (at which point the power in R1 is 512 mW). Because the basic input resistance of the AD8307 is not precise, VR1 is used to center the signal range at its input, doubling as a 4 dB intercept adjustment. The low frequency response extends to 15 Hz; a higher corner frequency can be selected as needed by scaling C1 and C2. The shunt capacitor C3 is used to lower the high frequency bandwidth to about 100 kHz, and thus lower the susceptibility to spurious signals. Other values should be chosen as needed for the coupling and filter capacitors.
Rev. C | Page 20 of 24
AD8307
1 W TO 1 kW 50 POWER METER
The front-end adaptation shown in Figure 41 provides the measurement of power being delivered from a transmitter final amplifier to an antenna. The range has been set to cover the power range -30 dBm (7.07 mV rms, or 1 W) to +60 dBm (223 V rms, or 1 kW). A nominal voltage attenuation ratio of 158:1 (44 dB) is used; thus the intercept is moved from -84 dBm to -40 dBm and the AD8307, scaled 0.25 V/decade of power, now reads 1.5 V for a power level of 100 mW, 2.0 V at 10 W and 2.5 V at 1 kW. The general expression is characteristic of log amps; indeed, the AD8307 exhibits the same scaling factor. The AD603 has a very low input referred noise: 1.3 nV/Hz at its 100 input, or 0.9 nV/Hz when matched to 50 , equivalent to 0.4 V rms, or -115 dBm, in a 200 kHz bandwidth. It is also capable of handling inputs in excess of 1.4 V rms, or +16 dBm. It is thus able to cope with a dynamic range of over 130 dB in this particular bandwidth. Now, if the gain control voltage for the X-AMP is derived from the output of the AD8307, the effect is to raise the gain of this frontend stage when the signal is small and lower it when it is large, but without altering the fundamental logarithmic nature of the response. This gain range is 40 dB, which, combined with the 90 dB range of the AD8307, again corresponds to a 130 dB range.
50 INPUT -105dBm TO +15dBm L1 750nH C1 150pF
P (dBm) = 40 (VOUT - 1)
The required attenuation could be implemented using a capacitive divider, providing a very low input capacitance, but it is difficult to ensure accurate values of small capacitors. A better approach is to use a resistive divider, taking the required precautions to minimize spurious coupling into the AD8307 by placing it in a shielded box, with the input resistor passing through a hole in this box, as indicated in Figure 41. The coupling capacitors shown here are suitable for f 10 MHz. A capacitor can be added across the input pins of the AD8307 to reduce the response to spurious HF signals, which, as already noted, extends to over 1 GHz. The mismatch caused by the loading of this resistor is trivial; only 0.05% of the power delivered to the load is absorbed by the measurement system, a maximum of 500 mW at 1 kW. The post-demodulation filtering and slope calibration arrangements are chosen from other applications described in this data sheet to meet the particular system requirements. The 1 nF capacitor lowers the risk of HF signals entering the AD8307 via the load.
TO ANTENNA 100k 1/2W 51pF NC VR1 2k INT 3dB 50 INPUT FROM P.A. 1W TO 1kW 604 51pF NC = NO CONNECT
8 7 6 5
R2 28k
0.65V
R1 187k
VP, +5V
BANDPASS FILTER*
4.7
0.1F NC
1 2 3 4
GPOS GNEG VINP
VPOS 8 VOUT 7 VNEG 6
R3 330 R4 464 VR1 5k INT 8dB
8
7
6
5
INP VPS ENB INT
AD603
AD8307
INM COM OFS OUT
1 2 3 4
COMM FDBK 5
NC 1nF R6 20k R7 80.6k
0.3V TO 2.3V
VN, -5V
R5 100k 0.15V TO 1.15V
*FOR EXAMPLE: MURATA SFE10.7MS2G-A NC = NO CONNECT
OUTPUT 10mV/dB
Figure 42. 120 dB Measurement System
0.1F
22
VP +5V
INP VPS ENB INT
AD8307
INM COM OFS OUT
1 2 3 4
LEADTHROUGH CAPACITORS, 1nF 2k OUTPUT
01082-041
NC 1nF
VOUT
Figure 42 shows how these two parts can work together to provide state-of-the-art IF measurements in applications such as spectrum/network analyzers and other high dynamic range instrumentation. To understand the operation, note first that the AD8307 is used to generate an output of about 0.3 V to 2.3 V. This 2 V span is divided by 2 in R5/R6/R7 to provide the 1 V span needed by the AD603 to vary its gain by 40 dB. Note that an increase in the positive voltage applied at GNEG (Pin 2 of AD603) lowers the gain. This feedback network is tapped to provide a convenient 10 mV/dB scaling at the output node, which can be buffered if necessary. The center of the voltage range fed back to the AD603 is 650 mV, and the 20 dB gain range is centered by R1/R2. Note that the intercept calibration of this system benefits from the use of a well regulated 5 V supply. To absorb the insertion loss of the filter and center the full dynamic range, the intercept is adjusted by varying the maximum gain of the AD603, using VR1. Figure 43 shows the AD8307 output over the range -120 dBm to +20 dBm and the deviation from an ideal logarithmic response. The dotted line shows the increase in the noise floor that results when the filter is omitted; the decibel difference is about 10 log10(50/0.2) or 24 dB, assuming a 50 MHz bandwidth from the AD603. An L-C filter can be used in place of the ceramic filter used in this example.
Figure 41. 1 W to 1 kW 50 Power Meter
MEASUREMENT SYSTEM WITH 120 dB DYNAMIC RANGE
The dynamic range of the AD8307 can be extended further-- from 90 dB to over 120 dB--by the addition of an X-AMP(R) such as the AD603. This type of variable gain amplifier exhibits a very exact exponential gain control characteristic, which is another way of stating that the gain varies by a constant number of decibels for a given change in the control voltage. For the AD603, this scaling factor is 40 dB/V, or 25 mV/dB. It is apparent that this property of a linear-in-dB response is
Rev. C | Page 21 of 24
01082-042
AD8307
2.50 2.25 2.00 1.75
VOUT (V)
WITHOUT FILTER
See Figure 40 for a more elaborate filter. To improve the law conformance at very low signal levels and at low frequencies, add C4 to the offset compensation loop.
2 1 0
ERROR (dB)
0.1F 5V 4.7
1.50 1.25 1.00 0.75 0.50 0.25 0 -100 -80 -60 -40 -20 INPUT LEVEL (dBm) 0 WITH FILTER ERROR (WITH FILTER)
-1 -2
VIN 0.5mV TO 20V SINE AMPLITUDE
C1 R1 10F 5k + C3 750pF
NC
8 7 6 5
INP VPS ENB INT
AD8307
INM COM OFS OUT
1 2 3 4
01082-043
20
Figure 43. Results for 120 dB Measurement System
NC = NO CONNECT
OPERATION AT LOW FREQUENCIES
The AD8307 provides excellent logarithmic conformance at signal frequencies that can be arbitrarily low, depending only on the values used for the input coupling capacitors. It can also be desirable to add a low-pass input filter in order to desensitize the log amp to HF signals. Figure 44 shows a simple arrangement, providing coupling with an attenuation of 20 dB; the intercept is shifted up by this attenuation, from -84 dBm to -64 dBm, and the input range is now 0.5 mV to 20 V (sine amplitude). A high-pass 3 dB corner frequency of nominally 3 Hz is set by the 10 F coupling capacitors C1 and C2, which are preferably tantalum electrolytics (note the polarity) and a low-pass 3 dB corner frequency of 200 kHz (set by C3 and the effective resistance at the input of 1 k). The -1% amplitude error points occur at 20 Hz and 30 kHz. These are readily altered to suit other applications by simple scaling. When C3 is zero, the low-pass corner is at 200 MHz. Note that the lower end of the dynamic range is improved by this capacitor, which essentially provides an HF short circuit at the input. This significantly lowers the wideband noise; the noise reduction is about 2 dB compared to when the AD8307 is driven from a 50 source. Ensure that the output is free of postdemodulation ripple by lowering the low-pass filter time constant. This is provided by C5; with the value shown in Figure 44, the output time constant is 125 ms.
+5V R1 4.7 C1 0.1F R2 3.3k VIN TEMP C1 1F
8 7 6 5
Figure 44. Connections for Low Frequency Operation
DC-COUPLED APPLICATIONS
It may occasionally be necessary to provide response to dc inputs. Since the AD8307 is internally dc-coupled, there is no fundamental reason why this is precluded. However, there is a practical constraint since its inputs must be positioned about 2 V above the COM potential for proper biasing of the first stage. If the source is a differential signal at this level, it can be directly connected to the input. For example, a microwave detector can be ac-coupled at its RF input and its baseband load then automatically provided by the floating RIN and CIN of the AD8307, at about VP/2. Usually, the source is a single sided ground-referenced signal; thus, it is necessary to provide a negative supply for the AD8307. This can be achieved as shown in Figure 45. The output is now referenced to this negative supply, and it is necessary to provide an output interface that performs a differential-to-single sided conversion. This is the purpose of the AD830. The slope can be arranged to be 20 mV/dB, when the output ideally runs from zero, for a dc input of 10 V, to 2.2 V, for an input of 4 V. The device is fundamentally insensitive to the sign of the input signal, but with this biasing scheme, the maximum negative input is constrained to about -1.5 V. The transfer function after trimming and with R7 = 0 is
VOUT = (0.4 V) log10 (VIN/10 V)
+5V FOR 20mV/dB +10V FOR 50mV/dB +15V FOR 100mV/dB -5V
8 7 6 5
VR2 50k R5*
VOUT
INP VPS ENB INT
VP X1
1
INT NC
VN Y2
4
AD8307
INM COM OFS OUT
1 2 3 4
AD830
X2
2
Y1
3
R7
20mV/dB AD589 R3 1k Q1 2N3904 -2V VR1 2k C3 0.1F R6 32.4k VR3 50k R9 250 R8
01082-044
-5V
NC = NO CONNECT
*51k FOR 20mV/dB; 5k FOR 100mV/dB
Figure 45. Connections for DC-Coupled Applications
Rev. C | Page 22 of 24
01082-045
+ R2 C2 10F 5k
C4 1F
C5 1F
VOUT 25mV/dB
AD8307
The intercept can be raised, for example, to 100 V, with the rationale that the dc precision does not warrant operation in the first decade (from 10 V to 100 V). Likewise, the slope can be raised to 50 mV/dB, using R7 = 3 k, R8 = 2 k , or to 100 mV/dB, to simplify decibel measurements on a DVM, using R7 = 8 k, R8 = 2 k, which raises the maximum output 11 V, thus requiring a 15 V supply for the AD830. The output can be made to swing in a negative direction by simply reversing Pins 1 and 2. Low-pass filtering capacitor, C3, sets the output rise time to about 1 ms.
6.0 5.5 5.0 4.5 4.0 1.0 0.5 0 -0.5 -1.0
Next, it is necessary to set the intercept. This is the purpose of VR2, which should be adjusted after VR1. The simplest method is to short the input and adjust VR2 for an output of 0.3 V, corresponding to the noise floor. For more exacting applications, a temporary sinusoidal test voltage of 1 mV in amplitude, at about 1 MHz, should be applied, which can require the use of a temporary on-board input attenuator. For 20 mV/dB scaling, a 10 V dc intercept (which is 6 dB below the ac intercept) requires adjusting the output to 0.68 V; for 100 mV/dB scaling, this becomes 3.4 V. If a 100 V intercept is preferred (usefully lowering the maximum output voltage), these become 0.28 V and 1.4 V, respectively. Finally, the slope must be adjusted. This can be performed by applying a low frequency square wave to the main input, having precisely determined upper and lower voltage levels, provided by a programmable waveform generator. A suitable choice is a 100 Hz square wave with levels of 10 mV and 1 V. The output is a low-pass filtered square wave, and its amplitude should be 0.8 V for 20 mV/dB scaling, or 4 V for 100 mV/dB scaling.
VOUT (V)
3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 10 100 1m 10m VIN 100m 1 10
ERROR (dB)
OPERATION ABOVE 500 MHZ
01082-046
Figure 46. Ideal Output and Law-Conformance Error for the DC-Coupled AD8307 at 50 mV/dB
Figure 46 shows the output and the law-conformance error, in the absence of noise and input offset, for the 50 mV/dB option. Note that the error ripple for dc excitation is about twice that for the more usual sinusoidal excitation. In practice, both the noise and the internal offset voltage degrade the accuracy in the first decade of the dynamic range. The latter is now manually nulled by VR1, using a simple method that ensures very low residual offsets. A temporary ac signal, typically a sine wave of 100 mV in amplitude at a frequency of about 100 Hz, is applied via the capacitor at node TEMP; this has the effect of disturbing the offset nulling voltage. The output voltage is then viewed on an oscilloscope and VR1 is adjusted until the peaks of the (frequency-doubled) waveform are exactly equal in amplitude. This procedure can provide an input null down to about 10 V. The temperature drift is very low, though not specified since the AD8307 is not principally designed to operate as a baseband log amp; in ac modes, this offset is nulled continuously and automatically.
The AD8307 is not intended for use above 500 MHz. However, it does provide useful performance at higher frequencies. Figure 47 shows a plot of the logarithmic output of the AD8307 for an input frequency of 900 MHz. The device shows good logarithmic conformance from -50 dBm to -10 dBm. There is a bump in the transfer function at -5 dBm, but if this is acceptable, the device is usable over a 60 dB dynamic range (-50 dBm to +10 dBm).
2.0 1.8 1.6 1.4
VOUT (V)
1.2 1.0 0.8 0.6 0.4 0.2 0 -60 -50 -40 -30 -20 PIN (dBm) -10 0 10
01082-047
Figure 47. Output vs. Input Level for a 900 MHz Input Signal
Rev. C | Page 23 of 24
AD8307 OUTLINE DIMENSIONS
0.400 (10.16) 0.365 (9.27) 0.355 (9.02)
8 1 5
4
0.280 (7.11) 0.250 (6.35) 0.240 (6.10)
0.100 (2.54) BSC 0.210 (5.33) MAX 0.150 (3.81) 0.130 (3.30) 0.115 (2.92) 0.022 (0.56) 0.018 (0.46) 0.014 (0.36) 0.070 (1.78) 0.060 (1.52) 0.045 (1.14)
0.325 (8.26) 0.310 (7.87) 0.300 (7.62) 0.060 (1.52) MAX 0.195 (4.95) 0.130 (3.30) 0.115 (2.92)
0.015 (0.38) MIN SEATING PLANE 0.005 (0.13) MIN
0.015 (0.38) GAUGE PLANE 0.430 (10.92) MAX
0.014 (0.36) 0.010 (0.25) 0.008 (0.20)
COMPLIANT TO JEDEC STANDARDS MS-001 CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN. CORNER LEADS MAY BE CONFIGURED AS WHOLE OR HALF LEADS.
Figure 48. 8-Lead Plastic Dual In-Line Package [PDIP] (N-8) Dimensions shown in inches and (millimeters)
5.00 (0.1968) 4.80 (0.1890)
4.00 (0.1574) 3.80 (0.1497)
8 1
5 4
6.20 (0.2440) 5.80 (0.2284)
1.27 (0.0500) BSC 0.25 (0.0098) 0.10 (0.0040) COPLANARITY 0.10 SEATING PLANE
1.75 (0.0688) 1.35 (0.0532)
0.50 (0.0196) 0.25 (0.0099) 8 0 0.25 (0.0098) 0.17 (0.0067) 1.27 (0.0500) 0.40 (0.0157)
45
0.51 (0.0201) 0.31 (0.0122)
COMPLIANT TO JEDEC STANDARDS MS-012-A A CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
Figure 49. 8-Lead Standard Small Outline Package [SOIC_N] Narrow Body (R-8) Dimensions shown in millimeters and (inches)
ORDERING GUIDE
Model AD8307AN AD8307ANZ1 AD8307AR AD8307AR-REEL AD8307AR-REEL7 AD8307ARZ1 AD8307ARZ-REEL1 AD8307ARZ-RL71
1
Temperature Range -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C
Package Description 8-Lead PDIP 8-Lead PDIP 8-Lead SOIC_N 8-Lead SOIC_N 13" REEL 8-Lead SOIC_N 7" REEL 8-Lead SOIC_N 8-Lead SOIC_N 13" REEL 8-Lead SOIC_N 7" REEL
060506-A
070606-A
Package Option N-8 N-8 R-8 R-8 R-8 R-8 R-8 R-8
Z = Pb-free part.
(c)2006 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D01082-0-10/06(C)
Rev. C | Page 24 of 24

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